A transmitter in a communication device such as a cellular phone or a wireless LAN (Local Area Network) is required to operate with a low (electric) power consumption while securing a desired output accuracy for transmitted signals, regardless of the degree of output power. In particular, a power amplifier circuit as a final step in the transmitter of the communication device is required to have a high power efficiency since the power amplifier circuit consumes 50% or higher of electric power consumed by the entire communication device.
In recent years, a switching amplifier circuit has attracted attention as a power amplifier circuit having a high power efficiency. The switching amplifier circuit simulates a pulse wave signal as an input signal and performs the power amplification while maintaining the relevant waveform. The pulse wave signal amplified by the switching amplifier circuit is emitted via an antenna toward the air after frequency components other than a desired frequency component are sufficiently reduced utilizing filter elements.
FIG. 12 shows an example of a circuit configuration for a class-D amplifier circuit as a representative of known switching amplification circuits.
The shown class-D amplifier circuit is formed by serially inserting two switch elements between a power supply and the ground. Complementary pulse signals as switching (opening and closing) control signals are input into the two switch elements so that only one of the switch elements is on (i.e., in the on position). When the switch element provided at the power supply is on and the switch element provided at the ground is off (i.e., in the off position), a voltage equal to the power supply voltage is output. In the reverse case, a voltage equal to the ground potential is output.
The above class-D amplifier circuit requires no bias current. Therefore, ideally, power loss is zero. In addition, the above switch elements can be implemented utilizing MOS (Metal-Oxide-Semiconductor) field-effect transistors or bipolar transistors.
FIG. 13 shows an example of a block configuration for a known transmitter that employs a class-D amplifier circuit (such a configuration may be analogized based on a signal generator in FIG. 1 of Non-Patent Document 1).
The transmitter has an RF signal generator, a driver amplifier, a class-D amplifier circuit, a filter, and the like, and amplifies an RF pulse signal (generated by the RF signal generator) utilizing the driver amplifier and the class-D amplifier circuit. A filter circuit next to the class-D amplifier circuit removes unnecessary components in the RF pulse signal amplified by the electric power so as to regenerate an RF radio signal.
The RF signal generator consists of a digital baseband processor, a polar converter, a ΔΣ (delta sigma) modulator, a comparator, and a mixer.
The polar converter converts orthogonal radio signals (I(t), Q(t)) generated by the digital baseband processor into an amplitude signal A(t) and a phase signal P(t) in accordance with the following formulas (1) and (2).[Formula 1]A(t)=√{square root over (I(t)2+Q(t)2)}{square root over (I(t)2+Q(t)2)}  (1)[Formula 2]P(t)=sin(ωc·t+α)  (2)
Here, α is computed by the following formula (3)[Formula 3]α=tan−1(I(t)/Q(t))  (3)
In addition, ωc denotes an angular frequency corresponding to a carrier frequency. Furthermore, as shown in the following formula (4), an RF radio signal RF(t) is a product of A(t) and P(t).[Formula 4]RF(t)=A(t)·P(t)  (4)
The comparator converts the phase signal P(t) having a sine waveform into a pulse signal having a rectangular form through a comparative operation utilizing a threshold of zero. The pulse phase signal PR(t) is represented by the following formula (5).[Formula 5]PR(t)=P(t)+H(t)  (5)
Here, H(t) is a harmonic component of P(t), that is generated when shaping P(t) to have a rectangular form.
The ΔΣ modulator operates in synchronism with a clock signal supplied from an external fixed clock source, and subjects the amplitude signal A(t) to ΔΣ modulation. As a specific example of the ΔΣ modulator, a configuration diagram of a known primary ΔΣ modulator is shown in FIG. 14. This ΔΣ modulator consists of delay devices, a quantizer, an adder, and a subtracter.
In this example, the quantizer is a 1-bit comparator that outputs a value of 1 or −1 by comparing the value of an input signal with a threshold. When the input signal and the output signal of the ΔΣ modulator are each denoted by Y(z) and W(z) and quantization noise generated in the quantizer is denoted by N(z), the following formula (6) is obtained between them.[Formula 6]W(z)=Y(z)+(1−z−1)·N(z)  (6)
In the above formula, z=ej(2πf/fs), where fs denotes a clock rate (i.e., sampling frequency) of the clock signal used in the ΔΣ modulator.
The above formula (6) means that the output signal W(z) includes the input signal and a component obtained by multiplying the quantization noise by (1−z−1) as a coefficient. The absolute value of (1−z−1) closes to zero in a frequency range that is sufficiently less than the sampling frequency, and the absolute value is 2 (maximum value) at a Nyquist frequency (that is defined as one-half of the sampling frequency).
When considering that a signal-to-quantization noise ratio here is represented by a ratio between Y(z) and (1−z−1)·N(z) in the above formula (6), in the present ΔΣ modulator, the lower the frequency range (i.e., sufficiently lower than the sampling frequency), the smaller the quantization noise, so that the noise is vanishingly low and a high signal-to-quantization noise ratio can be obtained. In contrast, the signal-to-quantization noise ratio is low in a relatively high frequency range.
That is, under a condition that the frequency band for signals is sufficiently low, the ΔΣ modulator can reduce incorporation of quantization noise into the relevant band. When A(z) denotes a signal obtained by subjecting the amplitude signal A(t) to Z conversion, the output signal of the ΔΣ modulator is represented by the following formula (7).[Formula 7]W(z)=A(z)+(1−z−1)·N(z)  (7)
When representing this formula in a time domain, the following formula (8) is obtained.[Formula 8]W(t)=A(t)+NH(t)  (8)
Here, NH(t) is the sum of a component obtained by representing the quantization noise (1−z−1)·N(z) (see above formula (7)) in a time region and an image component of A(t) that appears when representing A(z) in a time region, where the image component appears within a range greater than or equal to the Nyquist frequency.
The mixer in FIG. 13 outputs a product of a signal output from the ΔΣ modulator and a signal output from the comparator. A signal MIX(t) output from the mixer is represented by the following formula (9).[Formula 9]MIX(t)=A(t)·P(t)+A(t)·H(t)+P(t)·NH(t)+NH(t)·H(t)  (9)
The first term of formula (9) corresponds to the radio signal RF(t) represented by the above formula (4). This means that the RF signal generator can generate a pulse signal that includes a radio signal. The radio signal can be amplified to have a desired level by inputting the pulse signal via the driver amplifier into the class-D amplifier circuit.
Although the unnecessary components from the second term in formula (9) are also amplified, the filter next to the class-D amplifier circuit removes components out of the band of the filter.